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Texas Instruments Incorporated
Data Acquisition
How the voltage reference affects
ADC performance, Part 3
By Bonnie Baker,
Senior Applications Engineer,
and Miro Oljaca,
Senior Applications Engineer
This article is Part 3 of a three-part series that investigates
the design and performance of a voltage-reference system
for a successive-approximation-register (SAR) analog-to-
digital converter (ADC). Part 1 (see Reference 1) exam-
ined the ADC characteristics and specifications, with a
particular interest in the gain error and signal-to-noise
ratio, while assessing how the voltage reference impacts
the ADC transfer function and DC accuracy. Part 2 (see
Reference 2) examined the voltage-reference characteris-
tics, focusing on how the voltage-reference noise produces
the most error at the converter’s full-scale range. Part 2
concluded by presenting a design for a voltage-reference
circuit that is appropriate for 8- to 14-bit converters. This
article, Part 3, tackles the challenge of designing a voltage-
reference circuit that is appropriate for converters with
16+ bits. Part 3 examines methods of improving noise
filtering and of compensating for losses caused by the
improved filters.
Basics of reducing voltage-reference noise
As discussed in Part 2, the two sources of noise in the ref-
erence voltage are the internal output amplifier and the
bandgap. The voltage-reference circuit from Part 2 that
was configured with an 8- to 14-bit ADC can be used as a
starting point to continue the discussion. The size of the
least significant bit (LSB) of any converter in a 5-V system
is equal to 5 V/2
N
, where N is the number of converter
bits. The 8-bit LSB size in this environment is 19.5 mV,
and the 14-bit LSB size is 305 µV. The target value for
voltage-reference noise should be less than these LSB
values. The bandgap noise of the circuit from Part 2 was
reduced by adding an external capacitor to the output to
create a low-pass filter. This circuit’s output noise can be
further reduced by adding another capacitor as a passive
low-pass filter. Figure 1 shows an example of such a design,
which uses a voltage reference from the Texas Instruments
(TI) REF50xx family. In this design, the 1-µF capacitor
(C
1
) provides a minimal 21-dB noise reduction at the inter-
nal bandgap reference. C
2
, in combination with the open-
loop output resistance (R
O
) of the voltage reference’s
internal amplifier (see Reference 4), further reduces the
output noise of the reference at the V
REF_OUT
pin. In this
case, the equivalent series resistance (ESR) of the 10-µF
ceramic capacitor (C
2
) is equal to 200 mΩ.
Figure 1. Voltage-reference design appropriate
for 8- to 14-bit converter
V
IN
Voltage
Reference
REF50xx
Op Amp
–
R
O
10 k
Bandgap
Reference
V
REF_OUT
+
C
10 µF
(Ceramic)
2
1 k
TRIM
C
1
1 µF
V
REF_IN
D
OUT
AIN
ADC
5
Analog Applications Journal
4Q 2009
www.ti.com/aaj
High-Performance Analog Products
Data Acquisition
Texas Instruments Incorporated
Figure 2 shows a fast-Fourier-transform (FFT) plot of
the output signal of the circuit in Figure 1. Note that the
output-noise level peaks at around 9 kHz because of the
response of the circuit’s internal amplifier to the capacitive
load (C
2
). This peaking is the main contributor to the
overall measured noise. This output noise, measured with
an analog meter over a frequency range of up to 80 kHz, is
approximately 16.5 µV
RMS
. If the voltage-reference circuit
was connected to the input of an ADC, the measured
noise across a 65-kHz frequency range would be 138 µV
PP
.
This noise level makes the solution in Figure 1 adequate
for 8- to 14-bit converters.
Reducing voltage-reference noise for an ADC
with 16+ bits
Since the voltage-reference circuit in Figure 1 would
introduce too much noise into a converter with 16+ bits,
another low-pass filter can be added to further reduce the
reference’s output noise. This filter consists of a 10-kΩ
resistor (R
1
) and an additional capacitor (C
3
) as shown in
Figure 3. The corner frequency of this added RC filter,
1.59 Hz, will reduce broadband noise as well as noise at
extremely low frequencies.
Figure 2. FFT plot of V
REF_OUT
signal of circuit in Figure 1
–40
f=131.0720 kHz
S
–60
–80
Peaking at ~9 kHz
–100
–120
–140
–160
0
65
Frequency Spectrum
(
32768-Point FFT
) (kHz)
Figure 3. Voltage-reference circuit with R
1
and C
3
added as filters
V
IN
Voltage
Reference
REF50xx
Op Amp
R
10 k
1
–
R
O
10 k
Bandgap
Reference
V
REF_OUT
+
C
10 µF
2
C
10 µF
3
1 k
TRIM
C
1
1 µF
6
High-Performance Analog Products
www.ti.com/aaj
4Q 2009
Analog Applications Journal
Texas Instruments Incorporated
Data Acquisition
Figure 4. FFT plot of V
REF_OUT
signal of circuit with RC filter added
–40
f=131.0720 kHz
S
–60
–80
–100
–120
–140
–160
0
65
Frequency Spectrum
(
32768-Point FFT
) (kHz)
Figure 4 shows that the addition of R
1
and C
3
has a sig-
nificant effect on the output noise for this system. The
9-kHz noise peak is gone. With this signal response, the
output noise of the reference circuit in Figure 3 becomes
2.2 µV
RMS
or 15 µV
PP
, a reduction of nearly 90%. This
improvement brings the noise level so well under control
that the voltage-reference circuit is now
appropriate for ADC resolutions of up to
20 bits.
This is encouraging; however, pulling
current through R
1
from the ADC reference
pin will corrupt the conversion by intro-
ducing a voltage drop equivalent to the
average charge level from the reference pin
of the ADC. Consequently, the output of this
new circuit will not be able to adequately
drive the ADC’s voltage-reference input. To
accomplish this, a buffer will need to be
added to the low-pass filters.
Adding a buffer to the voltage-
reference circuit
Figure 5 shows an example of the fluctua-
tions in ADC reference drive current that
can occur during a conversion. The signal
was captured with a low-capacitance probe
to show the voltage drop across the 10-kΩ
resistor (R
1
) between the input of the ADC
voltage-reference pin and V
REF_OUT
. The top
trace in Figure 5 shows the trigger signal
that the converter receives to initiate a new
conversion. The ADC’s voltage-reference
circuit demands different amounts of current (bottom
trace) for the initiation of the conversion and for each
code decision. Therefore, the voltage-reference analog cir-
cuitry connected to the ADC must be able to accommo-
date these high-frequency fluctuations efficiently while
maintaining a strong voltage reference for the converter.
Figure 5. Drive current required by ADC’s reference input
Sample and
Convert Trigger
(
5 V/div
)
Drive Current for ADC’s
Voltage-
Reference Input
(
500 µA/div
)
Time
(
1 µs/div
)
7
Analog Applications Journal
4Q 2009
www.ti.com/aaj
High-Performance Analog Products
Data Acquisition
Texas Instruments Incorporated
Figure 6. Voltage-reference circuit with added buffer and output filter
V
IN
Voltage
Reference
REF50xx
Op Amp
OPA350
R
10 k
1
–
R
O
R
O_OPA350
–
10 k
Bandgap
Reference
V
REF_OUT
+
C
10 µF
C
10 µF
2
3
+
C
10 µF
4
1 k
TRIM
C
1
1 µF
V
REF_IN
D
OUT
AIN
ADC
OPA350’sopen-loopoutputresistance(R
O_OPA350
) modify
the open-loop voltage-gain (A
OL
) curve to create a margin-
ally stable state. To illustrate this phenomenon, Figure 7
shows how the output capacitor (C
4
), with a 0.2-Ω ESR
andtheOPA350’sopen-loopoutputresistance(43Ω),
modifiestheOPA350’sA
OL
curve. These curves can be
used to quickly determine the stability of the circuit. A
circuit with good stability would basically be one where
the rate of closure of the operational amplifier’s modified
A
OL
curve and closed-loop voltage-gain (A
CL
) curve is
Figure 6 shows a voltage-reference circuit that will ade-
quately drive a high-resolution ADC. In this circuit, the TI
OPA350isplacedasabufferafterthelow-passfilterthat
was constructed with R
1
and C
3
for the circuit in Figure 3.
TheOPA350drivesa10-µFfiltercapacitor(C
4
) and the
voltage-reference input pin of the ADC. The noise mea-
suredattheoutputoftheOPA350inFigure6is4.5µV
RMS
or 42 µV
PP
.TheinputbiascurrentoftheOPA350is10pA
at 25°C. This current, in combination with the current
through R
1
, generates a 100-nV, constant-DC drop. Note
that this voltage drop does not change with
the ADC’s bit decisions. It is true that the
inputbiascurrentoftheOPA350changes
over temperature, but a maximum current
that is no more than 10 nA at 125°C can be
expected. This value generates a change of
100 µV over a temperature range of 100°C.
It is useful to put the voltage drop across
R
1
into perspective. This voltage drop is added
to the errors contributed by the REF50xx
andtheOPA350.Theinitialerrorofthe
REF50xx output is ±0.05%, with an error
over temperature of 3 ppm/°C. With a 4.096-V
reference (REF5040), the initial reference
error is equal to 2.05 mV at room tempera-
ture and an additional 1.23 mV at 125°C.
Therefore, the reference output error is sig-
nificantly larger than the errors produced by
R
1
andvariationsintheOPA350’soffsetand
input bias current.
Amplifier stability
There is a final word of caution about the
circuit in Figure 6. The stability of the
OPA350canbecompromisedifC
4
and the
Figure 7. Frequency response of buffer with an RC load
140
120
100
Open-Loop Gain (A
)
f
pole
OL
80
Modified A
OL
60
40
20
f
zero
0
–20
Closed-Loop Gain (A
)
CL
–40
1
10
100
1 k 10 k 100 k
Frequency (Hz)
1 M
10 M
100 M
8
High-Performance Analog Products
www.ti.com/aaj
4Q 2009
Analog Applications Journal
Texas Instruments Incorporated
Data Acquisition
20 dB/decade. This rule of thumb is presented in
Reference 4. The open-loop output resistance of the
OPA350is43Ω, and the ESR of C
4
(R
ESR_C4
) is 200 mΩ.
The frequency locations of the pole and zero that are
created by these values are
Document Title
TI Lit. #
3. Bonnie Baker, “A Glossary of Analog-to-
Digital Specifications and Performance
Characteristics,” Application Report .........
sbaa147
4.TimGreen.Operationalamplifierstability,
Parts 3, 6, and 7.
EN-Genius Network:
analogZONE: acquisitionZONE
[Online].
Available: http://www.en-genius.net/includes/
files/acqt_
000000
.pdf (Replace “
000000
” with
“030705”
for Part 3,
“070405
” for Part 6, or
“052906”
for Part 7.)
1
f
=
××
=
368 Hz and
pole
2
π
(
R
+
R
)
×
C
OOPA
_
350
ESRC
_
4
4
1
f
=
××
= 79.6 kHz.
zero
2
π
R
×
C
—
ES
RRC
_4
4
5.BonnieC.BakerandMiroOljaca.(2007,
June 7). External components improve
SAR-ADC accuracy.
EDN
[Online].Available:
http://www.edn.com/contents/images/
6447231.pdf
Per Figure 7, the circuit in Figure 6 is stable.
Thinking ahead
Unfortunately,thevoltage-referencedesignsinthisarticle
can degrade ADC performance by adding unwanted tem-
perature drift and initial gain error. Higher-performance
systems with 21+ bits may require a voltage-reference
design that addresses these issues. Future articles will
explore a new approach with auto-zero amplifiers that will
compensate for these errors.
References
For more information related to this article, you can down-
load an Acrobat
®
Reader
®
file at www-s.ti.com/sc/techlit/
litnumber
and replace “
litnumber
” with the
TI Lit. #
for
the materials listed below.
Document Title
—
6.Wm.P.(Bill)Klein,MiroOljaca,andPete
Goad. (2007). Improved voltage reference
circuits maximize converter performance.
AnalogeLab™Webinar[Online].Available:
http://dataconverter.ti.com
(Scroll down to
“Videos” under “Analog eLab™ Design
Support” and select webinar title.)
—
7. Art Kay. Analysis and measurement of
intrinsic noise in op amp circuits, Part I.
EN-Genius Network: analogZONE:
audiovideoZONE
[Online].Available:
http://www.en-genius.net/includes/files/
avt_090406.pdf
TI Lit. #
—
1.BonnieBakerandMiroOljaca,“Howthe
Voltage Reference Affects ADC Performance,
Part 1,”
Analog Applications Journal
(2Q 2009) ...............................
slyt331
2.MiroOljacaandBonnieBaker,“Howthe
Voltage Reference Affects ADC Performance,
Part 2,”
Analog Applications Journal
(3Q 2009) ...............................
slyt339
Related Web sites
dataconverter.ti.com
www.ti.com/sc/device/OPA350
www.ti.com/sc/device/REF5040
9
Analog Applications Journal
4Q 2009
www.ti.com/aaj
High-Performance Analog Products
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